Conventionally, a receiver-side apparatus performs processing (hereinafter referred to as “frequency offset compensation) for compensating for a carrier frequency drift between a transmitter-side apparatus (for example, base station apparatus) and a receiver-side apparatus (for example, communication terminal apparatus).
The following will explain the radio receiving apparatus that performs a conventional frequency offset compensation with reference to FIG. 1, FIG. 2, and FIG. 3. FIG. 1 is a schematic view showing a state of a known symbol transmitted to the radio receiving apparatus that performs a frequency offset compensation. FIG. 2 is a block diagram showing a configuration of the radio receiving apparatus that performs a conventional frequency offset compensation. FIG. 3 is a schematic view showing a timing state of a known symbol of a path received by the radio receiving apparatus that performs a conventional frequency offset compensation.
The transmitter-side apparatus (not shown) transmits a signal including a known symbol 11 spread with Code A and a known symbol 12 spread with Code B. In this case, it is assumed that a length of Code A and a length of Code B are set to tCA and tCB, respectively, and that a distance between the known symbol 11 and known symbol 12 is set to tgap.
The signal transmitted from the transmitter-side apparatus is received via an antenna 21 by a radio receiving apparatus shown in FIG. 2. In FIG. 2, a signal received (received signal) by the antenna 21 is converted into a baseband signal from a carrier frequency signal by a reception RF section 22. At this time, a local signal sent from a crystal oscillator 38 (to be described later) is used at the reception RF section 22. An in-phase component (I-ch) of the baseband signal and a quadrature phase component (Q-ch) thereof are output to an A/D converter 23 and an A/D converter 24 from the reception RF section 22, respectively.
The baseband signal with I-ch and the baseband signal with Q-ch are converted into digital signals by the A/converter 23 and A/D converter 24, respectively. The baseband signal with I-ch and the baseband signal with Q-ch, which are converted into digital signals, are output to a searcher 25, a despreader 26, and a despreader 27.
The searcher 25 examines the correlation between the baseband signal converted to the digital signal and Code A, which is the known code, to detect a code timing (namely, timing of each path) with which power of a correlation value reaches a maximum value as illustrated in FIG. 3. The searcher 25 also detects timing of Code B using the detected code timing. For example, if a timing difference between path 1 of Code A and path 2 thereof is set to tp, timing of code B of path 1 becomes tA+tgap, and timing of Code B of path 2 becomes tA+Tgap+tp. Thus, timing of Code B is also calculated based on the detected timing of Code A. In this way, despread timing at the despreaders 26 and 27, pilot timing at a channel estimating section 28 and path timing at a RAKE combining section 29 are calculated by the searcher 25.
Timing of Code A and Code B of path 1 is output to the despreader 26 from the searcher 25, and timing of Code A and Code B of path 2 is output to the despreader 27 from the searcher 25. Timing of Code A and Code B of path 1, and timing of Code A and Code B of path 2 are output to the channel estimating section 28 from the searcher 25. Moreover, timing of path 1 and timing of path 2 are output to the RAKE combining section 29 from the searcher 25.
The despreader 26 provides despread processing using Code A and Code B to the baseband signal with I-ch based on timing of Code A and Code B of path 1 from the searcher 25. Similarly, the despreader 26 provides despread processing to the baseband signal with Q-ch using Code A and Code B based on timing of Code A and Code B of path 1 from the searcher 25, respectively. Moreover, the despreader 26 provides despread processing to the baseband signals with I-ch and Q-ch using a predetermined spreading code (spreading code assigned to the present radio receiving apparatus). The baseband signals with I-ch and Q-ch subjected to despread processing are output to the channel estimating section 28 and RAKE combining section 29.
The despreader 27 provides despread processing using Code A and Code B to the baseband signal with I-ch based on timing of Code A and Code B of path 2 from the searcher 25. Similarly, the despreader 27 provides despread processing to the baseband signal with Q-ch using Code A and Code B based on timing of Code A and Code B of path 2 from the searcher 25, respectively. Moreover, the despreader 27 provides despread processing to the baseband signals with I-ch and Q-ch using a predetermined spreading code. The baseband signals with I-ch and Q-ch subjected to despread processing are output to the channel estimating section 28 and RAKE combining section 29.
The channel estimating section 28 extracts a signal, which corresponds to the known symbol 11 and known symbol 12, from among baseband signals with I-ch and Q-ch subjected to spread processing from the despreader 26 based on timing of Code A and Code B of path 1 from the searcher 25. A channel estimation value of path 1 is calculated using this extracted signal. Likewise, the channel estimating section 28 extracts a signal, which corresponds to the known symbol 11 and known symbol 12, from among baseband signals with I-ch and Q-ch subjected to spread processing from the despreader 27 based on timing of Code A and Code B of path 2 from the searcher 25. A channel estimation value of path 2 is calculated using this extracted signal. The channel estimation values of path 1 and path 2 calculated by the channel estimating section 28 are output to the RAKE combining section 29.
The RAKE combining section 29 multiplies the baseband signal with I-ch and Q-ch subjected to despread processing from the despreader 26 by an inverse characteristic of the channel estimation value of path 1 from the channel estimating section 28. The RAKE combining section 29 multiplies the baseband signal with I-ch and Q-ch subjected to despread processing from the despreader 27 by an inverse characteristic of the channel estimation value of path 2 from the channel estimating section 28. Moreover, the RAKE combining section 29 RAKE combines the despread baseband signal with I-ch and Q-ch of path 1 multiplied by the inverse characteristic of channel estimation value with the despread baseband signal with I-ch and Q-ch of path 2 multiplied by the inverse characteristic of channel estimation value based on timing of path 1 and path 2 from the searcher 25.
The baseband signal with I-ch and Q-ch subjected to RAKE combining is output to a modulating section 30. The modulating section 30 provides demodulation processing to the baseband signal with I-ch and Q-ch subjected to RAKE combining, whereby obtaining received data.
The baseband signal with I-ch subjected to RAKE combining is output to a complex correlation calculating section 33. Also, after the baseband signal with I-ch subjected to RAKE combining is delayed by tAB (=tCA/2+tgap+tCB/2; see FIG. 1) by a delay section 31, and the resultant is output to the complex correlation calculating section 33. Similarly, the baseband signal with Q-ch subjected to RAKE combining is output to a complex correlation calculating section 33. Likewise, after the baseband signal with Q-ch subjected to RAKE combining is delayed by tAB by a delay section 32, and the resultant is output to the complex correlation calculating section 33.
The complex correlation calculating section 33 performs complex correlation processing using the baseband signal with I-ch subjected to RAKE combining from the RAKE combining 29 and the baseband signal with I-ch subjected to RAKE combining and delayed by tAB from the delay section 31. Moreover, the complex correlation calculating section 33 performs complex correlation processing using the baseband signal with Q-ch subjected to RAKE combining from the RAKE combining 29 and the baseband signal with Q-ch subjected to RAKE combining and delayed by tAB from the delay section 32. The signals with I-ch and Q-ch subjected to complex correlation processing are output to a phase estimating section 34.
The phase estimating section 34 calculates a phase rotation amount per unit time using the signals with I-ch and Q-ch, which are subjected to complex correlation processing and which are sent from the complex correlation calculating section 33. A smoothing section 35 calculates a frequency offset using the phase rotation amount calculated by the phase calculating section 34. The calculated frequency offset is output to a control voltage converting section 36.
The voltage converting section 36 converts the calculated frequency offset into a control voltage to the crystal oscillator 38. This control voltage is converted into an analog signal by a D/A converter 37, and the resultant is output to the crystal oscillator 38. In this way, the frequency of local signal is controlled at the crystal oscillator 38. The frequency offset compensation is thus carried out.
However, the conventional radio receiving apparatus that performs the frequency offset compensation has the following problem. Namely, in the conventional radio receiving apparatus that performs the frequency offset compensation, since the phase rotation amount is estimated using the baseband signals subjected to RAKE combining, there is a problem in which accuracy of phase rotation amount to be estimated is decreased particularly when Dopplar frequency caused by high-speed moving becomes high.
For example, as illustrated in FIG. 4, a channel estimation value is calculated using a known symbol placed at the central portion of the slot. In a case where the use of this channel estimation value is shared in the slot, accuracy of the channel estimation value deteriorates with distance from the channel estimation segment, so that accuracy of the baseband signals subjected to RAKE combining deteriorates. As a result, accuracy of phase rotation amount to be estimated is decreased. In other words, accuracy of phase rotation amount to be estimated depends on accuracy of channel estimation using the baseband signals subjected to RAKE combining.
The factors that decrease accuracy of the phase rotation amount to be estimated will be explained with reference to FIGS. 5A, 5B, 5C, 5D, 5E, 5F, 6A, 6B, and 6C.
FIG. 5A is a schematic view showing a state of the phase rotation amount of the baseband signal obtained by despread processing using Code A of path 1. FIG. 5B is a schematic view showing a state of the phase rotation amount of the baseband signal obtained by despread processing using Code B of path 1.
As illustrated in FIGS. 5A and 5B, since channel estimation of path 1 is carried out on a path-by-path basis by the channel estimating section 28, a channel estimation value obtained using Code A and a channel estimation value obtained using Code B are substantially the same as each other (Δθ1ch).
The baseband signal obtained by despread processing using Code A (hereinafter simply referred to as “baseband signal of Code A”) rotates against a transmit signal by a phase variation (Δθ1fad) due to fading. The baseband signal obtained by despread processing using Code B (hereinafter simply referred to as “baseband signal of Code B”) rotates against the baseband signal of Code A by (66 θAB).
FIG. 5C is a schematic view showing a state of a phase rotation amount of the baseband signal obtained by despread processing using Code A of path 2. FIG. 5D is a schematic view showing a state of a phase rotation amount of the baseband signal obtained by despread processing using Code B of path 2.
Because of a difference between path 1 and path 2 in the propagation path, the baseband signal of Code A rotates against the baseband signal of Code A of path 1 by a phase rotation amount (Δθp) and a phase variation (Δθ2fad) due to fading. In addition, the phase rotation amount (Δθp) is a phase rotation amount corresponding to a time difference (tp) between path 1 and path 2. The baseband signal of Code B further rotates against the baseband signal of Code A of path 1 by ΔθAB.
Next, attention will be paid on the baseband signal subjected to RAKE combining by the RAKE combining section 29. FIG. 5E is a schematic view showing a state of a phase rotation amount of the baseband signal of Code A subjected to RAKE combining. FIG. 5F is a schematic view showing a state of a phase rotation amount of the baseband signal of Code B subjected to RAKE combining.
As illustrated in FIG. 5E, the baseband signal subjected to RAKE combining of Code A (namely, the baseband signal, which is obtained by RAKE combining the baseband signal of Code A of path 1 and the baseband signal of Code A of path 2) becomes a signal including a channel estimation error (Δθch. errA).
Likewise, as illustrated in FIG. 5F, the baseband signal subjected to RAKE combining of Code B (namely, the baseband signal, which is obtained by RAKE combining the baseband signal of Code B of path 1 and the baseband signal of Code B of path 2) becomes a signal including a channel estimation error (Δθch. errB) and a phase rotation amount (ΔθAB) due to a frequency offset to be calculated.
FIG. 6A is a schematic view showing a state of the channel estimation error of the baseband signal of Code A subjected to RAKE combining. FIG. 6B is a schematic view showing a state of the channel estimation error of the baseband signal of Code B subjected to RAKE combining. FIG. 6C is a schematic view showing a state of a signal to be subjected to complex correlation processing at the conventional radio receiving apparatus that performs frequency offset compensation.
In other words, as illustrated in FIGS. 6A, 6B, and 6C, since complex correlation processing is performed using the baseband signals of Code A and Code B each which includes the channel estimation error and which is subjected to RAKE combining, the signal obtained by this complex correlation processing includes the channel estimation error. Accordingly, the finally obtained phase rotation amount due to the frequency offset includes a channel estimation error, that is, an error corresponding to (Δθch. errA+Δ θ ch. errB). As a result, at the high-speed moving time at which channel estimation accuracy deteriorates, the estimation error of the phase rotation amount due to particularly the frequency offset deteriorates, resulting in a decline in the quality of a demodulated signal.